Marc and Yuhang

For DDS1, we found the signals used for SHG/IRMC/OPO demodulation are smaller than 7dBm. However, the mixers require at least 7dBm signals. Therefore, we decide to install amplifiers for corresponding DDS channels. After the installation, the situation of DDS1 is shown as follows:

DDS1 CH0: DDS output = -8dBm output (for SHG/IRMC modulation)

DDS1 CH1: DDS output+18dB amplifier+power splitter = 7dBm output *2 (for SHG/IRMC demodulation)

DDS1 CH2: DDS output =  -8dBm output (for OPO modulation)

DDS1 CH3: DDS output+18dB amplifier= 10dBm output  (for OPO demodulation)

Since we are using internal amplifiers for SHG/IRMC/OPO demodulation, their old amplifiers will not be used.

We did the test and found that DDS1 board works well and they are outputing good values. Then we optimized all PDH signals, measured TFs and error signals. The results of measurements are attached.

Marc and Yuhang

The first test of DDS3 board showed problem about USB connection. Therefore, we checked the connection and soldering of USB. We found a soldering problem related to USB connector. After that, we tried to solder it again. Following problems about soldering cost us quite a lot of time:

1. The wires going through the breadboard holes are not straight, which make wires very diffcult to be removed

2. A relatively large hole needs to be soldered in order to fix USB on the breadboard. When the solder is applied to an inappropriate side of this large hole, it causes the USB outer shell touching ground. We checked a working DDS board, whose USB's outer shell is not connected to ground. Due to this inappropriate solder, we wasted a USB connector.

3. When fixing USB on the breadboard, we need to choose the two sides of the breadboard. But only one side will make a correct connection.

After solving these problems, we tested DDS3 board output signal magnitude and put attenuator to get required level of signals.

DDS3 CH0: DDS output+18dB amplifier+12dB attenuator = -2dBm output (for PLL CC)

DDS3 CH1: DDS output+18dB amplifier+12dB attenuator = -2dBm output (for PLL ppol)

DDS3 CH2: DDS output+18dB amplifier+power splitter = 7dBm output *2  (for CC1 and CCFC)

DDS3 CH3: DDS output+18dB amplifier+10dB attenuator = 0dBm output  (for CC2)

We compared the PLL phase noise for the cases of using -8dBm LO and -2dBm LO. From the datasheet of ADF4002, it requires LO from -5dBm to 2dBm. Therefore, we should prefer -2dBm LO. Figure 1 and 2 show the comparison of PLL phase noise. However, the shape of phase noise curve is not in agree with the measurement done in elog863, which needs further investigation.

On the other hand, higher LO also makes a higher phase noise. This is out of our expectation.

Michael, Marc and Yuhang

DDS signals usually give an output of -6dBm, which is not enough for many mixers. Due to the lack of enough LO power, we had issues, such as CCFC error demodulation. To solve this problem, Matteo ordered several amplifiers. The idea is to put them inside the DDS board and connect the DDS output directly to them.

Yesterday, Aso-san kindly provided us an instruction before the implementation of these amplifiers. Today, we followed the design of Matteo and implemented part of those amplifiers (for DDS2 and DDS3).

Figure 1 shows the connection done for an amplifier (We did five in total for today).

Figure 2 shows the DDS2 board before putting amplifier (we found unfiltered CH1 output is giving signal).

Figure 3 shows the DDS2 board after putting the amplifier.

Then I took it to TAMA and did several tests. In the beginning, I found the signal was not present in CH1. Then I changed CH1 from unfiltered CH1 to filtered CH1(shown in attached figure 4). After this, I discovered that signal (shown in figure 5) increase from -8dBm to 9dBm after amplifier implementation. This signal is used as LO to demodulate the filter cavity length error signal for GR. Figures 6 and 7 show the check of PDH amplitude for these two cases. The PDH becomes a bit smaller with a larger LO. I compared TF and GR locking length noise with these two cases.

Figure 8 shows TFs. After implementing the amplifier, the unity gain frequency is smaller while the phase margin is better. The amplified case also shows a better phase for higher (compared with UGF) frequency region.

Figure 9 shows error signals. After implementing the amplifier, the integrated length noise becomes less. This error signal is not calibrated. Besides, it maybe better to compare them again when they have almost the same unity gain frequency.

All amplifiers are also installed inside DDS3. We will test it tomorrow.

The time (around JST 2am 21st Dec 2020) of this sudden change has coincidence with an earthquake.

Marc and Yuhang

We found difficulty to align FC on this Monday. Then we checked oplev signals and found a sudden position change for END mirror (figure 1, we didn't find sudden change on INPUT).

By changing DC offset for END, its oplev sensing signals were recovered (figure 2).

Marc, Michael and Yuhang

In september, we had problem of FDS measurement, which is the FDS feature (rotation from sqz(asqz) to asqz(sqz)) disappeared when we locked CCFC loop. After that, it has been long time we didn't measure FDS again. Recently, we would like to change the FDS configuration. So we decide to redo the FDS measurement before going on. The important degradation sources are as following:

Green power: 25mW (11dB generated squeezing)

optical losses: 68%

CC2 loop introduced a 30dB attenuator for error signal (We found CC2 loop always had oscillation & CC2 loop could be only closed when the gain is small)

other parameters are kept the same with setting in this Feb. Besides, detuning was chosen to be 114Hz (this number is calculated from entry 2296)

The measurement was sometimes not stable today. This instability was the squeezing level going up and down for each measurement. However, this happened while CC1 and CC2 loop were both locked(a bit strange for me, we could check those spectrums). Anyway, we found some stable moments and took measurement.

According to the mentioned degradation sources, I tried to fit data and got results in attached figure 1. The detuning was fit to a larger detuning compared with setting. Besides, the fit curves don't match well with data. However, the good point is that FDS could be measured again with CCFC loop closed. 

As mixers need to be operated in saturation mode, I temporarily take the amplifier channel used for IRMC demodulation to amplify the signal from DDS. DDS provides about -6dBm signals. With the mentioned amplifier, LO was amplified to about 5dBm. At the same time, the RF signal was about -15dBm.

When we scan the CCFC phase with a sine wave, the demodulated signals will deviate from sine wave if the demodulation process has problems. So I did this test with 5dBm LO (shown in attached figure 1) for different RF power (-15dBm, -9dBm, -6dBm and -3dBm). These tests are in attached figures 2 to 5. All these figures seem to provide good shape demodulated signals (sinusoidal). From these figures, we could also see that the pk-pk signal also increases with the increase of RF power almost linearly (115mV, 206mV, 288mV, 380mV).

I also checked the CCFC error signals for these cases(figure 6,7,8). They are consistent with the error signals we found in elog2308. And apparently, better SNR is achieved with -3dBm RF power.

(We could add 12dB+12dB+3dB attenuator for the -3dBm signal to simulate a factor of ~25 decreases of CCSB power)

I fitted the measured CCFC error signal by fitting the CC detuning (with respect to carrier), demodulation phase, and starting time (first plot). In this plot, misalignment effect is not considered.

In the second plot, I added the misalignment effect in theoretical curve by fixing the mode matching to 94%.

Matteo and Yuhang

The suppression of filter cavity length noise provides stable detuning, which is vital for the production of frequency dependent squeezing. The CCFC control loop is designed to achieve this goal.

To understand better how CCFC control works, several characterization works have been done recently. They are listed as follows:

1. Figure 1 shows many length error signals and noise curves. The addition of CCFC error signal introduces length noise for GR loop at low frequency. This is validated by figure 3 and 4. The GR+IR error signal doesn't change because the filter cavity length change doesn't change.

2. Figure 2 shows correction signals. For the correction signals send to the main laser or end mirror, they are the same whether there is CCFC or not. This is consistent with the unchange of IR+GR error signal.

3. Figures 3 and 4 show FC GR TRA/REF DC spectrums. CCFC causes the GR length noise increase, which translates into intensity noise. 

In elog2231 and elog2267, a worse locking accuracy was found to be caused by AA.

Today I compared the FC_IR_TRA while AA is on or off. It seems AA induced noise increase doesn't have the same shape with FC length noise (but similar).

This noise increase is clearly visible but could be well suppressed if CCFC lock is implemented.

12dB attenuator was added for RF signal (before the 32dB amplifier)

12dB attenuation was applied to LO signal (DAC current control was reduced from 1/2(-12dBm) to 1/8(-24dBm))

Current RF amp: -15dBm

Current LO amp: -24dBm

[Aritomi, Yuhang, Matteo]

We found that we still had saturation problem of CCFC RF and LO so we reduced them.

Then we measured CCFC error signal with different CCFC demodulation phase (Pic. 1). AOM FM freq is 300 mHz and deviation is 2kHz, so AOM scan speed for IR is 4kHz/(5/3 s)/2 = 1.2 kHz/s. CCFC amplitude for normalization is 28 mV. The calibration factor of CCFC error signal is determined by fitting the blue curve around 0, which is 1191 Hz/V.

We locked CCFC with 70 deg and 250 deg CCFC demodulation phase (both are I phase, but sign is opposite) and compared the locking accuracy with CCFC lock (Pic. 2). We found that CCFC locking accuracy with 250 deg is smaller than 70 deg above 1kHz. Changing CCFC demod by 180 deg means that CCSB on resonance and off resonance are swapped. CCSB noise on resonance is filtered out by cavity pole while the noise of other CCSB is not. If noise of upper/lower CCSB are different, this noise difference can happen.

IR filter is 500 gain and 30 Hz low pass filter.

Anyway now CCFC locking accuracy is below 1Hz if the calibration factor is correct. Strange thing is that locking accuracy above 10kHz is much better than BAB locking accuracy with green lock.

Elog2300 described optimization for CCFC error signal. To characterize better these error signals, I put measured CCFC error signal as follows.

Figure 1 is CCFC error signal at different demodulation phase, after modematching optimization.

Figure 2 is CCFC error signal at different demodulation phase, after mixer optimization.

Matteo and Yuhang

As reported in elog2300, we optimized mode matching and mixer. We obtained a larger CCFC error signal after that. Then we used it to lock the filter cavity length for IR. Control loop information is summarized as follows:

1. Gain of CCFC loop: 50
2. Corner frequency of CCFC loop: 30Hz (one order low pass)
3. Error signal shape: figure 1
4. Open loop transfer function (only CCFC part): figure 2 (40mVpk-pk excitation used)
5. Error signal spectrum (loop on/off): figure 3.
6. Calibration for error signal: AOM speed (4000Hz/2.5s)(figure 4) divided by error signal slope around zero (75mV/43.7ms)(figure 1 and 5) divided by 2 (AOM scan green to IR) : 4000*43.7/2.5/75/2 = 466 Hz/V

We could see that the CCFC method stabilized length noise for IR below ~1kHz. The IR length noise reached 2.3Hz after closing the CCFC loop. Compared with the old case, we could see that the main difference in IR length noise is around 1kHz~10kHz. The reason for this difference is still unknown. But if the CCFC error signal can go back to the old case, the CCFC loop can reduce IR length noise to less than 1Hz.

By changing CCFC demodulation phase, CCFC error signal offset should change in a sinusoidal way. I checked this after the optimization of mixer. The result is in attached figure 1.

I checked the histogram of CCFC and CC1 error signal. This check is after the mixer optimization.

We could see that strange behavior of histogram reported in elog2302 disappeared.

Matteo and Yuhang

As reported in elog2289 and elog2302, the demodulated signal from mixer ZX05-1L-S+ has strange behaviors, such as not exactly sinusoidal or strange data distribution. We realized these issues but we didn't know what is the reason.

On 2020/22/07, we checked two quadrature-phase signals of CCFC error signal while CC1 phase is scanned more than 2pi. While checking, we found these two quadrature-phase signals were not the same. Attached figure 1 shows these two signals.From this figure, the quadrature-phase signal is quite similar with sinusoidal shape while the in-phase one is quite linear between each maximum and minimum. After observing this difference, we start to investigate what is the difference between these two channels.

Comparison of these two channels:

1. The RF signals come from the same PD, the LOs come from the same channel of DDS3

2. LO signal is splitted by ZMSCQ-2-90, RF signal is splitted by ZFDC-10-1-S+

3. They use the same mixer ZX05-1L-S+ and the same low pass filter SLP-1.9+

The splitting of LO makes one LO ~11dB smaller than the other one (The splitting of LO should give identical output. However, there is difference due to frequency issue.). The splitting of RF makes one RF ~10dB smaller than the other one. (RF signal is about -3dBm before splitting)

In the end, we found the problem comes from LO. We were using ~-6dBm LO, which is smaller than the datasheet requirement. However, in practice, this mixer needs even smaller LO (-12dBm LO is used now).

When CC1 is locked and the filter cavity is detuned, the CCFC error signal only shows an offset. This field should be identical with CC1 error signal if offset is not considered. Before the optimization of the mixer, we checked the histogram of this offset. From the attached figure, we could see that this histogram has some problems (no data located in the center). It could come from an oscillation of this signal.

We should recheck it after the optimization of the mixer. 

The filter cavity GR lock's OLTF may differ with the filter cavity GR+IR lock's OLTF at low frequency. Therefore, we start to investigate GR OLTF's low-frequency part.

In the attached figure, there are four measurements. Their legends are listed in the sequence of time on 2020/12/07. We could see that:

1. All measurement shows flat gain at low frequency, which is different from what we expect.

2. Morning and evening measurements' magnitude are quite different at low frequency. The reason for this difference is still unknown.

3. Measurement phases are different with/without SR560 (just passing through without gain/filters). We could see that the phase margin is better if SR560 is used.

Matteo and Yuhang

Based on Aritomi-san's code, I add the degradation from mode-matching to the CCFC error signal. The simulation result is in attached figure 1. From this simulation, worse mode-matching makes CCFC error signal degrade around resonance. But mode-matching doesn't affect the CCFC error signal's offset.

Based on this simulation, we sent BAB to the filter cavity and checked the mode-matching was about 0.75. We found the IR drift happened only in the yaw direction. After optimizing yaw, mode matching increased to about 0.9. When we checked the CCFC error signal's pk-pk value, we found some issues with this signal's demodulation. After optimizing the mixer, we saw an even better CCFC error signal. The comparison of CCFC error signals before and after optimization is in attached figure 2.

I compared the mm-optimized/mm-original CCFC error signal's minimum. In the simulation, the ratio is 0.64. While in measurement, it is 0.58.